Getting the Voids Out   Mar. 23, 2020

Getting the Voids Out Image.png

Reducing solder voids is a pervasive challenge affecting the electronics industry.  With higher thermal dissipation requirements placed upon the attachment of a device to a circuit board the number of voids within the solder joint becomes critical.  There are various considerations that need to be taken into account, including 1) the design of the substrate 2) the surface condition of the part being attached 3) the type of solder selected 4) the reflow profile 5) pattern and volume of paste deposition, among others.  Let’s dive further into some of these considerations:

Substrate design can have a large effect on the resulting level of solder voiding within the solder joint connection.   Most land pads use thermal vias to dissipate heat while parts requiring even higher thermal dissipation capability utilize embedded heat slugs.  These thermal dissipation structures can impact how the solder connection is formed during reflow, causing temperature differences in areas that have vias or slugs vs. areas that do not.

Solder paste selection and flux type can also impact the final overall solder void connection.  In general, for bottom terminated components (BTC) a ‘no clean flux’ within the paste should be used.   That said, many solder paste manufacturers have their own formulations to help with reducing the amount of solder voiding.   Careful attention should be paid to selecting the right solder paste/flux formulation and to follow the supplier’s recommended solder profile.   In addition, the Association Connecting Electronics Industries (IPC) has a standard IPC-7530A: Guidelines for Temperature Profiling for Mass Soldering Processes to assist with understanding the reflow process and setting up reflow profiles.

The solder paste application process can also play a role in how many solder voids are formed.  In most cases stencil printing is performed.  There are various methods to print the paste in terms of pattern to allow flux volatiles to escape.    Balancing the volume of paste deposited on the I/Os vs. the ground paddles on the mating substrate, can also impact the amount of voiding.         

MACOM application note S2083 is available to customers to assist in the recommended land pad design and stencil design for our products.  In addition, for BTCs, IPC also has available document IPC-7093: Design and Assembly Process Implementation for Bottom Termination (BTC) Components to help customers with the overall construction and post assembly of BTC components onto a second level assembly. 


For more information and supplemental reading:

1. M. Johnson, E. Eilenberg, P. Hogan, J. Aldrich, A. Reyes, “Comparison of Laminate Construction Methods on Fabrication, Junction Temperature and Second Level Assembly”, 2019 IPC Apex Expo, January 2019, San Diego, CA United States.

2. M. Johnson, “Techniques to Reduce Solder Voiding Under BTC Components”, 2018 IMAPs New England, May 2018, Boxborough, MA United States.



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Non-Linear Transmission Line Comb Generators Part-2: Solutions to the Low Phase Noise Problem   Jan. 21, 2020

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In this two part series from MACOM, we will delve into Non-Linear Transmission Line (NLTL) Comb Generators.  In part 1 we considered the phase noise problem and introduced a potential solution to the problem. In this second part of the blog series, we will explore NLTL comb generation, compare it to its predecessor comb generation using Step Recovery Diodes and see how the NLTL comb generation approach can enable improved sensitivity and lower bit error rates in communication systems.

SRD Comb Generation

Many systems require signals at frequencies which are not easily generated or in some cases impossible to generate directly.  The widely-employed mitigation strategy has been to apply a locally-generated, low-frequency signal known as the ‘fundamental frequency’ to a circuit containing nonlinear impedance, which translates energy from the fundamental signal frequency to its harmonics.  Step recovery diodes (SRDs) have been used extensively to multiply the frequency of a signal.  The step recovery diode is a pn junction device, typically silicon, comprising 3 layers: the p layer anode, a lightly-doped n layer and a heavily-doped n layer, the latter two of which comprise the cathode of the diode.

The SRD goes into conduction when the positive (forward bias) alternation of an input signal is applied.  Charge carriers (holes) from the p layer and the n layers (electrons) flow, which produces low diode impedance.  Immediately after the polarity of the input signal reverses to the negative-bias polarity, the population of these charge carriers is as large as it was under forward bias, so the diode’s impedance remains low.  A non-zero interval is required for the charge carriers to be conducted out of the diode. 

When the population of the free charge carriers is small, the recombination of holes and electrons becomes the dominant mechanism responsible for the impedance of the diode.  When recombination is completed, the diode’s impedance has completed the transition from its low value to its maximum value.  In effect, the diode current “snaps off”. In Figure 1 below, the typical current versus time plot is shown.

Figure 1.jpg

Figure 1 Typical Current vs. Time Plot


A typical step-recovery-diode frequency multiplier circuit is shown in Figure 2


Figure 2 Typical Step-Recovery Diode Frequency Multiplier Circuit


The RF current which flows through the SRD, D, also flows through the inductor Li.  As the SRD current snaps off, the current through inductor Li also very rapidly decreases to zero.  This sudden decrease in current through the inductor creates an impulse-like voltage waveform which is rich in even and odd harmonics.

Recombination is a stochastic process.  Recombination can only occur when a hole and an electron pair are in “the right place at the right time”.  It is not difficult to imagine that this process does not occur identically, cycle after input cycle, but rather occurs with random variations.  This produces jitter of the impulse-like waveform in the time domain, which is equivalent to phase noise in the frequency domain.


NLTL Comb Generation

MACOM’s family of nonlinear transmission line (NLTL) comb generators produce harmonics of an input signal in an entirely different manner than that employed in the SRD comb generator.

Recall that the equivalent circuit of a transmission line is composed of a ladder network of series inductors and shunt capacitors.  This structure is shown below.


In a nonlinear transmission line comb generator, the shunt capacitors are replaced with Schottky junction varactor diodes.  The capacitance of a varactor diode is inversely proportional to the reverse-bias voltage across the diode.  At small reverse-bias signal voltage, the varactor diode produces maximum capacitance.  As the reverse voltage increases in magnitude, the capacitance of the varactors decreases in a nonlinear manner. 



The capacitance of the Schottky varactors is a virtually instantaneous function of the amplitude of an incident signal.  As an input signal propagates through the transmission line structure, the phase velocity of the lower-voltage portion of its negative alternation is lower than that of its higher-voltage portion due to the higher varactor capacitance produced by the Schottky varactors.(3)  Cascaded L-C sections of the transmission structure cause a portion of the waveform to be increasingly vertical, as plotted versus time, as additional L-C sections are traversed.  This distortion of the waveform produces harmonics of the fundamental input signal frequency.

Schottky varactor diodes are utilized in the NLTL comb generator because they are majority carrier devices - there are no minority charge carriers in Schottky diodes so there is no stochastic carrier recombination to produce flicker noise and other random fluctuations which are unavoidable in pn junction devices like SRDs.  Thus an NLTL comb generator produces much less additive phase noise than does a SRD comb generator, by as much as 10 to 15 dB.  Figure 3 below compares the additive phase noise for offset frequencies up to 1 MHz for the MLPNC-7103S1-SMT580 NLTL comb generator at 12 GHz output frequency when driven with a 500 MHz input signal at 22 dBm input power versus the output of a SRD comb generator with at the same output frequency with the same input signal conditions.



Figure 3


MACOM offers a family of low-phase-noise NLTL comb generators in surface mount or connectorized modules which have best-in-class phase noise, conversion loss and frequency coverage performance.  These products enable significantly improved sensitivity in radar and communications receivers and lower bit error rates in vector-modulated systems.  For more information on MACOM’s Comb Generators:



3. Breitbarth, J., “Design and Characterization of Low Phase Noise Microwave Circuits”, University of Colorado, 2006

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Non-Linear Transmission Line Comb Generators Part-1: The Phase Noise Problem and Comb Generation   Jan. 07, 2020

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In this two-part series from MACOM, we will delve into Non-Linear Transmission Line (NLTL) Comb Generators, first understanding the phase noise problem, and understanding a potential solution to the problem. In the second part of the blog series, we will explore NLTL comb generation, compare it to its predecessor comb generation using Step Recovery Diodes and see how the NLTL comb generation approach can enable improved sensitivity and lower bit error rates in communication systems.

The Phase Noise Problem

Let’s start with the problem with circuits requiring low noise performance. Below we see a block diagram of the RF and IF portions of a typical superheterodyne receiver.  A weak signal is received at the antenna – 1) optionally amplified by a low noise amplifier, 2) filtered to reduce the effects of broadband noise and interferer signals whose frequencies may be close to that of the desired signal and then 3) downconverted to a lower, intermediate frequency for further processing. 


Figure 1: Typical Superheterodyne Receiver

In the ideal case, the downconverter mixer mixes the received signal with a single-frequency local oscillator signal.  In the real case though, the local oscillator signal never comprises a single frequency, but is always accompanied by close-in noise sidebands which are generated in the local oscillator signal chain.  Also, the received signal may be accompanied by close-in interfering signals which cannot be completely removed by the band pass filter.     

What are the Noise Sidebands? – Phase and Amplitude Noise

This noise is composed of phase noise and amplitude noise.  Phase noise is random, instantaneous variations in the phase of a signal, in this case, the local oscillator signal.  Amplitude noise may be thought of as random, instantaneous variations in the amplitude of the signal.

The net effect of the noise sidebands is to self-jam very weak received signals whose frequencies are close to the local oscillator frequency.  Because the noise sidebands are so close to the local oscillator frequency that they cannot be removed by filtering, the internally-generated phase and amplitude noise must be minimized in any receiver intended to process weak signals or signals accompanied by interferer signals.

In addition to the self-jamming problem, in communications system which utilize phase modulation, phase noise can also cause high bit error rates.  Regardless of the type of modulation utilized, phase noise can cause spectral regrowth which can produce adjacent channel interference(1) and reciprocal mixing which can mask a low-amplitude received signal(2).

A Solution? Low Phase Noise Signal Generation

Low-phase-noise signal generation can be accomplished using several design approaches, such as crystal oscillators, ceramic resonator oscillators and SAW/BAW resonator oscillators.  One factor which is common to these various types of circuits is that they are only capable of producing signals at relatively low frequencies (typically in the HF up to the mid-UHF bands) due to the electromechanical modes of operation of their resonators. 

Comb Generation

Many systems require signals at frequencies which are not easily generated or in some cases impossible to generate directly.  The widely-employed mitigation strategy has been to apply a locally-generated, low-frequency signal known as the ‘fundamental frequency’ to a circuit containing nonlinear impedance, which translates energy from the fundamental signal frequency to its harmonics.  Such circuits are known as comb generators since these harmonics, when depicted in the frequency domain, resemble a comb.

Tune in for Part 2 in this series for more information on NLTL Comb Generators. Members of MACOM’s applications engineering team are ready to help you select the optimal diodes and circuit topologies for your application. For more information on MACOM’s solutions, visit:


(1). Khanzadi, M. R., “Phase Noise in Communications Systems”, Chalmers University of Technology, November 2015

(2). Henkes, D. D., “Analyzing the Role of Local Oscillator Phase Noise in Reciprocal Mixing”, Microwave Products Digest, October 2013

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Designing with Diodes: Protecting Sensitive Components   Apr. 29, 2019

receiver protect limiter blog.jpg (!blank)Sensitive low noise amplifiers (LNAs) in radar or radio receivers cannot tolerate large input signals without sustaining damage. What’s the solution? Receiver-protector limiter (RPL) circuits, the “heart” of which typically comprises PIN diodes, can be utilized to protect sensitive components from large input signals without adversely affecting small-signal operation.

RPL circuits do not require external control signals. These circuits comprise at least one PIN diode connected in shunt with the signal path, along with one or more passive components, such as RF choke inductors and DC-blocking capacitors. A simple (but possibly complete) RPL circuit is shown below.

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When there is no RF input signal or when only a small RF input signal is present, the impedance of the limiter PIN diode is at its maximum value, the magnitude of which is typically in the few-hundreds of ohms or greater. Consequently, the diode produces a very small impedance mismatch and correspondingly low insertion loss.

When a large input signal is present, the RF voltage forces charge carriers into the PIN diode’s I layer, holes from its P layer and electrons from its N layer. The population of free charge carriers introduced into the I layer lowers its RF resistance, which produces an impedance mismatch as seen from the RPL circuit’s RF ports.

This mismatch causes energy from the input signal to be reflected to its source. The reflected signal, in concert with the incident signal, produces a standing wave with a voltage minimum at the PIN diode since it temporarily presents the lowest impedance along the transmission line. There is a current maximum collocated with every voltage minimum along the transmission line. This current flows through the PIN diode, enhancing the population of free charge carriers in the diode’s I layer, which results in lower series resistance, a greater impedance mismatch and a “deeper” voltage minimum. Eventually the diode’s resistance will reach its minimum value, which is determined by the design of the PIN diode and the magnitude of the RF signal. Increases in the RF signal amplitude force the diode into heavier conduction, thus further reducing the diode’s resistance until the diode is saturated and produces its lowest possible resistance. This results in an output power vs. input power curve as shown below.

graph 2.PNG

After the large RF signal is no longer present, the diode’s resistance remains low (and its insertion loss remains large) if the population of free charge carriers in the I layer is large. Upon cessation of the large RF signal, the population of free charge carriers will decrease by two mechanisms: conduction out of the I layer and recombination within the I layer. The magnitude of the conduction is determined primarily by the DC resistance in the current path external to the diode.

The rate of recombination is determined by several factors, including the free-charge-carrier density in the I layer, the concentration of dopant atoms and other charge-trapping sites in the I layer, etc. Due to the required parameters of the diodes, the greater the RF signal which a PIN diode can safely handle, the longer its recovery time to low insertion loss will be.

The properties of the I layer of the PIN diode determine how this circuit performs. The I layer’s thickness (sometimes referred to as its width) determines the input power at which the diode goes into limiting – the thicker the I layer, the higher the input-referred 1 dB compression level (also known as the threshold level). The thickness of the I layer, the area of the diode’s junction and the material of which the diode is made determine the resistance of the diode as well as its capacitance. These parameters also determine the diode’s thermal resistance.

The simplest implementation of a PIN RPL circuit comprises a PIN diode, an RF choke inductor and a pair of DC blocking capacitors. The RF choke inductor is critical to the performance of the RPL circuit, with the primary function to complete the DC current path for the PIN diode. When a large signal forces charge carriers into the diode’s I layer, a DC current is established in the diode. If a compete path for this DC current is not provided, the diode’s resistance cannot be reduced, and no limiting can occur. This current will flow in the same direction as a rectified current would flow, but it is not produced by rectification.

Implementation of the choke inductor in the RPL circuit can be challenging, since inductors are the least ideal of the components in the RPL circuit. Inductors all have series and parallel resonances due to their inductance and their parasitic inter-winding capacitance. Care must be taken to ensure that series resonances do not occur within the operating frequency band. Additionally, the choke’s DC resistance must be minimized in order to reduce the recovery time of the RPL circuit.

Note: the DC blocking capacitors are optional. They are only necessary if there are DC voltages or currents present on the input or output transmission lines which might bias the PIN diode.

A Practical Example

Assume the maximum input power which an LNA can tolerate is 15 dBm. This power level sets the requirement for the I layer thickness of the PIN diode in the RPL circuit, which in this case is approximately 2 microns. A designer can then determine the acceptable capacitance of the PIN diode from the frequency of the RF signal and the maximum acceptable small-signal insertion loss. If they assume the RPL operates in X Band and the maximum acceptable insertion loss is 0.5 dB, then the maximum capacitance of the diode can be calculated.

The insertion loss (IL) in decibels of a shunt capacitance is given by:

equation 1.PNG

We can solve that equation for C:

equation 2.PNG

For f = 12 GHz, IL = 0.5 dB and Z0 = 50 Ω, C = 0.185 pF.

Along with the I layer thickness, this value of capacitance will determine the area of the diode’s junction.

The combination of thin I layer and small junction area creates a diode which has relatively high thermal resistance, which cannot dissipate very much power without forcing the junction temperature to exceed its maximum rated value of 175 °C. Typically, a 2 micron diode with 0.185 pF capacitance can safely handle a large CW input signal of around 30 to 33 dBm. A larger signal can potentially damage or immediately destroy this diode due to the Joule heating produced by the current flowing through the diode’s electrical resistance.

PIN diode RPL circuits reliably protect sensitive components like low noise amplifiers in radar or radio receivers from large incident signals. For RPL applications which require very low flat leakage output power but high input power handling, additional diode stages and other circuit enhancements can be added at the input side of the RPL circuit.

Members of MACOM’s applications engineering team are ready to help you select the optimal diodes and circuit topologies for your RPL application. For more information on MACOM’s solutions, visit:

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